Cmos image sensor for direct time of flight measurement

ABSTRACT

A direct TOF optic sensor is based on CMOS pixels, wherein a pixel structure comprises a photodetector PhD, a non linear resistance R and a transfer MOS transistor in series, and delivers an output signal at a sensing node SN between the resistor and the transfer transistors. The photogenerated current is continuously drained into the nonlinear resistance and converted to a voltage signal by the RC circuit formed by the nonlinear resistance and a capacitance at the sense node SN. The voltage signal is continuously transmitted to a readout circuitry  300  having a fast analog to digital converter. The RC circuit within the pixel structure has a low pass filtering function and a high frequency integrating function, so that noise, in particular thermal noise due to the nonlinear resistance is mainly shifted in a low frequency range, separate from a high frequency range of the main signal component corresponding to a pulse light signal received at the photodetector. The main signal component is recovered by means of one of a band pass or high pass filter F implemented in the readout circuitry, that increases the signal to noise ratio in the high frequency range.

FIELD OF THE INVENTION

The invention relates generally to optical sensors for time-of-flight(TOF) measurement, and concerns more particularly optical sensors basedon CMOS imaging sensor, and direct time of flight measurement (D-TOF).

BACKGROUND

Time of flight optical sensors are used in 3D imaging and range-findingapplications, for example for survey or automotive driving assistanceapplications, to provide a map of distances of any object/detail in ascene under observation. Each distance information is computed from atime measure by a pixel or a group of pixels in the optical sensor,which is the travel timet_(t) of a pulse light emitted from a modulatedlight source close to the detector towards the scene, and backscatteredby an object in the illuminated scene. The applicable equationisd=(½C)·t_(t), where C is the light speed, and the division factor by 2is in account of the round trip travel of light. In one method, the timemeasure is computed indirectly, from phase differences. In this case,the light source is a RF modulated one and the imaging sensor isoperated to measure a light amplitude captured in each of N integrationperiods (N phases) that are offset to one another in a capturingsequence and the travel time is derived from the N levels using wellknown equations. In another method, the travel time is measured directlythrough detection of rising edges of pulse signals in a reconstructedwaveform. There, the light source is one that emits light pulses and theimage sensor is operated to trigger rising edges of backscattered shortlight pulse signals.

In any case, the optical sensor must have a high dynamic range and goodsignal to noise ratio (SNR), to be able to detect the nearest andfarthest objects in the scene and/or darkest and brightest ones: thismeans that the pixel structure and operation should bring in the lowestnoise level for enabling detection of the weakest signals overbackground light noise, and should not saturate for the strongestsignals. The constraint may vary according to the intended application.But for instance, in automotive driving assistance applications, theoptical sensor should be efficient when used by night, to capture anydark and bright details (same by daylight when the car enters a tunnelor a dark passage). Or even when a strong ambient light exists, like thesun light. CMOS image sensors are good candidates for such applications,with efficient pixels in terms of quantum efficiency, fill factor andhigh dynamic range. Also, the readout method with correlated doublesampling (CDS) is efficient to cancel thermal noise (so called kTCnoise) being brought by the electronic (reset transistor) in thecapacitive sense node of the CMOS pixel. But in practice, CMOS imagesensors in TOF systems implement only the indirect method, and theirpixels are then operated in a very conventional timing sequencecomprising a reset phase, an integration (N phases) sequence, and then areadout sequence.

However there is a growing interest in D-TOF systems implementing thedirect measurement. The general principle of a D-TOF system isschematically illustrated on FIG. 1. It comprises a light pulse source10 and an image sensor 11 close to one another, and a timer circuit 12comprising at least one counter (processing system). A time count isinitiated in the timer circuit by a start-count signal at the time ofemission of a light pulse LP_(E) by the light source 10 towards a 3Dscene in a field of view; and the time count is stopped by a stop-countsignal on detection by the image sensor 11 of a rising edge of abackscattered light pulse LP. Many time counts may be used,corresponding to objects at different distances and/or different regionsof interest in the matrix of pixels of the sensor. This is well knownstate-of-the-art.

As known D-TOF systems we can cite SPADs systems (where “SPADs” meansSingle-Photon Avalanche Diodes). These systems are very efficient andaccurate but needs a lot of power, because they rely on statisticalanalysis, histograms and averaging to find the rising edges in thesampled waveform, which implies a lot of light pulses. The efficiency isbecause of the great sensitivity of SPADs, to a single photon, and theinsensitivity to readout noise. However their fill factor is quitesmall, because each pixel must include the photodetector (avalanchephotodiode) together with a quenching circuit, a time to digitalconverter and an histogram module; and their quantum efficiency is alsosmall (probability that one photon reaching the photodetector generatesone electron), which combined with the fill factor gives a poorphotodetection efficiency (PDE): only a limited number of photons amongthe overall incident photons can be detected in each pixel, let us sayonly one every twenty photons or even more.

There is then a need for high dynamic ranging D-TOF systems able toprovide an accurate response with no use of statistical tools, based onthe emission of only a few number of short light pulses, for instance ina range of one to five light pulses at most, and preferably based ononly one or two pulses, minimizing both the need in light source powerand circuit occupation area. All this is an incentive trend to look atdesigning D-TOF systems based on CMOS image sensors in which the pixelstructure is comprised of a photodetector (a photodiode, preferably apinned photodiode, or a photogate), a capacitive sensing node (floatingdiffusion) and a number of transistors (or gates of transistors) for thesequencing of the initialization, integration and readout phases of thepixel.

For D-TOF operation however, the CMOS pixels cannot be conventionallyoperated with a separate integration phase, then readout phase, but in away that enables continuous reconstruction (readout) of the currentwaveform signal photogenerated by the sensitive element of the pixel.

A basic idea to achieve this is to sample the signal generated at thephotosensitive node at very high frequency (Nyquist-Shannon theorem)with respect to the pulse duration so that the waveform of a singlepulse of signal can be reconstructed (as an oscilloscope does), whichallows to minimize the light power and have a very short reaction time.However the signal originated from the photodetector is a current flow,and it is very difficult to send out a very weak current flow to beconverted to digital directly in current without be strongly degraded bythe noise.

In more details, for D-TOF operation, the signal information to triggerin the continuous readout waveform is a rising edge corresponding to thetime information to measure (time position of the backscattered lightpulse). The rising edge physically corresponds to the photogeneration ofa few electrons over the short pulse duration of a backscattered lightpulse incident on the pixel. This may correspond in practice to a veryweak signal amounting to a few electrons only, 10 to 20 electrons forexample, which means that the internal noise at the pixel level must bevery low. There is another requirement which is to be able to detectsuccessive backscattered light pulses that can be close in time. Let'stake a practical numerical example to better understand the underlyingproblem: with the light pulses having a 5 nanoseconds pulse duration,the receiver should be able to distinguish among two close backscatteredlight pulses corresponding to different details in the scene that areseparated by only some tens of nanoseconds, for instance 20 nanoseconds.This means that at the sense node, which is capacitive, the memoryeffect of the rising edge must be short, to avoid mixing of pulses,which means loss of information.

It is then proposed to achieve this through integrating thephotogenerated current produced by the photodetector in the capacitivesensing node of the pixel, and using a leakage resistor to discharge thecapacitor and make it ready to integrate a next pulse. However the RCcircuit acts as a high frequency integrator which modifies the shape ofa pulse signal. Further, a high resistor value may be desirable toincrease the gain of the I/V conversion for weak signals, but then forstrong signals (that would correspond to bright and close details in theilluminated scene) the sense node capacitance may saturate, which meansloss of information. Another constraint is linked to the RC timeconstant, which should not be too high so that after a step voltage atthe sense node as a result of the high frequency integration by the RCcircuit receiving an incoming pulse of current, the step can vanishfastly enough to enable detection of a new incoming pulse of current.

Another concern is the thermal noise (kTC noise) automatically broughtby the resistor in the capacitive sense node of the CMOS pixel. If CDSreadout is efficient for cancelling the thermal noise in theconventional chronological sequence, it is not a compatible technic withthe proposed continuous signal reconstruction sequence, because of thehigh sampling rate which is needed. This thermal noise adds to the shotnoise which directly depends on the signal level, which means the noiseis not constant in time.

Then, operating a CMOS pixel for high dynamic range D-TOF measurementappears not straightforward at least in view of these different issues,in particular high dynamic range with practical constraints regardingsaturation and SNR, and shape recovering, and noise and signal analysisat the level of the pixel appears necessary to understand respectiveinfluence of R and C over signal and noise. This analysis has inparticular highlighted that a key point regarding SNR and influence ofthermal noise, is that thermal noise power depends only of the capacitorvalue, not that of the resistance as assumed at first glance. Further,we found that the rms value of the thermal noise, when itis expressed interms of a voltage value, is equal to the square root of the ratio ofthe product of the Boltzmann constant k_(B) by the absolute temperatureT to the capacitance value. Then it comes that the rms voltage isreduced with a high capacitance value.

However, we found that when seeking at distinguishing a weak signal ofonly a few electrons corresponding to a single short light pulse asexplained above, over noise, we should better express the rms value notin terms of a voltage, but of a number of electrons. Then we found thatto reduce the influence of thermal noise, the capacitance value shouldin fact be reduced as much as possible. Let us briefly explain this withreference to a simplified model in small signals of a CMOS imaging pixelimplementing continuous I/V conversion of the photocurrent I(t) which isillustrated on FIG. 2. The RC circuit is in parallel on the currentsource I(t), which represents the photogenerated current. For the noiseanalysis, which depends on the signal, it was assumed that the currentsource emits a pulse of amplitude I₀ and of duration τ, (where τ is theduration of the short light pulse of the D-TOF system of FIG. 1). Thecapacitor represents the integration capacitance at the sense node SN ofthe pixel. The resistor in parallel allows to make the signalinformation vanish in a certain time (RC time constant) and be ready toreceive a new incoming current pulse.

For the purpose of evaluating the effect on SNR at the pixel level (shotnoise, thermal noise) let us take a realistic example of a weak signalof N_(e−)=16 electrons produced from a light pulse of 5 nanoseconds(pulse duration). First, shot noise: an average (rms) value of the shotnoise intrinsic to the signal then amounts to 4 electrons in applicationof EQ1: σ_(shot) _(Ne−) =√{square root over (N_(e−))} and the signal toshot noise ratio, which is the ratio

$\left( \frac{{Ne} -}{\sqrt{N_{e -}}} \right),$

amounts to 4 electrons, which is correct (it is generally admitted thatthe minimum SNR is 3 electrons).

Second, thermal noise: an average (rms) voltage value of the thermalnoise induced by the resistor R is obtained from equation EQ2:

${{\sigma_{R_{\upsilon}}\sqrt{\overset{\_}{\upsilon^{2}}}} = \sqrt{\frac{k_{B} \cdot T}{C}}},$

where k_(B) is the Boltzmann constant, T the absolute temperature (300°K) and C the capacitance value.

Expressed in terms of electrons, it gives:

$\begin{matrix}{\sigma_{R_{{Ne} -}} = {\frac{\sigma_{R_{\upsilon}} \cdot C}{q} = \frac{\sqrt{k_{B} \cdot T \cdot C}}{q}}} & ({EQ3})\end{matrix}$

where q is the electric charge of one electron.

This shows as indicated above, that the thermal noise in terms ofelectrons increases with the capacitance value. Then to minimize thermalnoise, we should consider a pixel structure and operation in which thecapacitance at the sense node is reduced as much as possible.

But this also demonstrates that even with a low capacitance value, whichtypically means lower than a few units of femto farads (10⁻¹⁵ farads)with now-a-days technology, the thermal noise remains too high, aboveten electrons. For instance with a capacitance value as low as 2fF, therms average value of the thermal noise amounts to 18 electrons (inapplication of EQ. 3). Then with a weak signal evaluated as above to 16electrons, the SNR taking in account both shot noise and kTC noise fallsto a value less than 1:

$\begin{matrix}{{{SNR} = {\frac{N_{e -}}{\sqrt{N_{e -} + \sigma_{R_{{Ne} -}}^{2}}} = {\frac{16}{\sqrt{{16} + {18^{2}}}} = 0}}},87} & ({EQ4})\end{matrix}$

This is far too small for successfully distinguishing a weak signal fromnoise. Note that in addition, there are further noise sources, not evendiscussed here, in particular background light noise, which depends onthe application (light operational conditions and kind of sensed scene)and from the electronics coming after the pixel (amplifier, sampling,A/D conversion).

Then, even if we can find R and C values to optimize the differentconstraints regarding shape reconstruction, memory effect, I/V gain andSNR, this will not be enough to efficiently enable detection of suchweak signals at the receiver based on continuous waveformreconstruction.

This low SNR together with the other constraints regarding pulse shapemodification and saturation limits are technical problems to solve forachieving an efficient D-TOF CMOS pixel.

SUMMARY OF THE INVENTION

In the invention we find a technical solution to this problem, whichcombines use of a nonlinear resistor inside the pixel with filteringtechnics in the readout circuit. A nonlinear resistor provides theadvantage that when the signal increases to a certain level, because ofa strong light signal, a strong light background or a too longmeasurement phase, the resistance value is reduced which limits furtherexcursion. That is, the nonlinear resistance avoids saturation in veryhigh dynamic range, which is desirable. But the nonlinear resistance wasfound to produce an interesting advantage when considering the RCfunction with regards to the frequency domain.

In more details, if we ignore the noise aspect, a backscattered lightpulse reaching the pixel results in the generation at the photodetectornode of a number of electrons over the light pulse duration σ, which isthe current pulse illustrated by way of example on FIG. 3, and resultsthrough continuous sensing by a nonlinear resistance combining with thesense node capacitance as proposed, in a fast rising of the voltagewaveform (steep edge) at the sense node as shown on FIG. 4.

This is because, the proposed sensing RC circuit does not function as alow pass filter over the full spectral range, but only in a lowfrequency range, and functions as an integrator over the capacitancenode in a high frequency range which is that of the main informationsignal (corresponding to the pulse rising edge) that is of interest. Wecould demonstrate this through asymptotic analysis of the transferfunction of the RC circuit of FIG. 2:

-   -   for

${f ⪡ \frac{1}{2 \cdot \pi \cdot R \cdot C}},$

corresponding to low frequencies, then the transfer equation is V=I·R;

-   -   for

${f\mspace{11mu}\text{⪢}\frac{1}{2 \cdot \pi \cdot R \cdot C}},$

corresponding to high frequencies, then the transfer equation is

${V = \frac{I}{s \cdot C}},$

which represents an integrator of the intensity over the capacitor:

this is exactly the same behavior that would be obtained if no resistorwas used at all.

This high frequency integration over the capacitance is what happens tothe main signal component corresponding to the (short) light pulseduration σ. For instance, with σ=5 ns, the frequency signal is equal to200 MHz: the useful signal information is then found above, for instancein a bandwidth of few tens of MHz above 200 MHz, and is not filtered bythe RC circuit in account of the practical resistance and capacitancevalues. However the shape of the signal is strongly degraded because ofthe RC time constant as shown on FIG. 4.

In contrast, as shown on FIG. 5, the thermal noise which is equivalentto white noise (flat spectral density), is concentrated in a lowfrequency band delimited by a cut-off frequency f_(lp) of the low passfilter created by the RC circuit (f_(lp)=1/(2πRC)).

In practice an upper limit of the low frequency range concentrating mostof the noise component corresponds to a cut-off frequency of the RCcircuit; and a lower limit of the high frequency range concentratingmost of the useful signal is a function of the pulse duration a, whichresults in practice in spaced apart (separated) low and high frequencybands.

Then, we ingeniously exploit this spectral separation resulting from thecontinuous current sensing through the proposed RC circuit inside thepixel, to increase the signal to noise ratio in the high frequency rangeof the useful signal through filtering, which also enables to recoverthe initial pulse shape. Then efficient detection of the pulse locationsis obtained through triggering on the digitized samples (afterfiltering).

The invention is then about a CMOS pixel structure for achieving directtime of flight measurement and proposes implementing a continuous I/Vconversion inside the pixel through a nonlinear resistor, and thetechnical solution takes advantage of a low pass filter operation andhigh frequency integration function of an RC circuit created by thenonlinear resistor combined with the sense node capacitance to recovermost of the information of the signal with high signal to noise ratiothrough filtering in a band pass or high pass range at the level of thereadout circuitry.

Other technical aspects improve the technical solution, through usingtransistors conventionally present in basic CMOS pixels to implement theI/V conversion in the measurement phase which is optimal in terms ofconception and fabrication costs, and efficient in terms of SNR, highdynamic range, and simplicity of operation.

In particular a transfer transistor is usually provided in a CMOS pixelbetween the photodetector node and the sense node for transferring thephotogenerated electrons accumulated in the photodetector to the sensenode, so that the readout operation of the pixel can start. In theinvention, such a transfer transistor between a photodiode and a sensenode is provided in the pixel for use as a decoupling element betweenthe photosensitive node and the sense node all along the measurementphase and the readout can be done while the transfer transistor isactive. The capacitance at the sense node is then minimized, which helpsto reduce the thermal noise as has been explained above.

As claimed, the invention concerns then a CMOS imaging sensor fordetecting time occurrence of light pulses having a given pulse durationτ, comprising:

-   -   pixels, having a pixel structure comprising at least:        -   a photodetector operating as a current source,        -   a transfer transistor connected in series between a sense            node and the photodetector,        -   a nonlinear resistor connected between a voltage supplying            node of the pixel and the sense node, the sense node being a            capacitive sense node,    -   a control circuit for controlling a measurement phase in at        least a selected pixel for measuring a time occurrence of a        light pulse reaching the photodetector in the pixel, wherein in        the selected pixel a voltage reference is applied to the voltage        reference node, and the transfer transistor is the ON state all        along the measurement phase, the transfer transistor then        operating as a decoupling element between a photodetector node        and the sense node, and the nonlinear resistor combining with a        capacitance at the sense node form a RC circuit that has a low        pass filtering function and high frequency integration function        that produce a voltage signal that has a main signal component        comprising at least a pulse location information in a high        frequency range, and a noise component mainly concentrated in a        low frequency range spaced apart from said high frequency range,        and    -   a readout circuitry of a voltage signal from the sense node of a        selected pixel, which comprises at least:        -   an analog to digital converter applying a high sampling time            in respect of the pulse width duration, and        -   filtering means configured to apply one of a band pass or            high pass filter, before or after analog to digital            conversion, having the effect of increasing the signal to            noise ratio in at least a frequency band around the main            signal components.    -   Implementing a non-linear resistor as a discrete resistor is in        practice expensive in area occupation.

Advantageously, the nonlinear resistor is implemented through atransistor operated in a sub-threshold region. In an embodiment, thistransistor is the reset transistor generally found in a conventionalCMOS pixel. According to the invention, this reset transistor isoperated in the sub-threshold region as a nonlinear resistor in selectedpixel(s), and used as a switch and turned on to maintain the sense nodeat a voltage reference in non-selected pixels.

The invention also concerns a direct time of flight system comprisingsuch an optical sensor as a receiver; and a method to operate a CMOSpixel for a measure of time through triggering a pulse location in avoltage signal waveform corresponding to a light pulse having reachedthe pixel.

Other characteristics and advantages of the invention will now bedescribed, by way of non-limiting examples and embodiments, withreference to the accompanying drawings, in which:

FIG. 1 illustrates a general principle of a D-TOF system;

FIG. 2 is an equivalent circuit in small signal of a CMOS pixelintegrating continuous I/V conversion through a resistor according tothe invention;

FIG. 3 represents an ideal pulse of current generated by thephotodetector of a CMOS pixel in response to a short (backscattered)light pulse;

FIG. 4 illustrates the RC high frequency integration function over thecapacitance of the sense node of the pulse of current of FIG. 3;

FIG. 5 represents the RC low pass filtering function applying to thethermal noise induced at the sense node capacitance by the resistor;

FIG. 6 is a schematic diagram of a basic pixel structure with associatedcircuits including an amplifier circuit between a sense node of thepixel and the input of a readout circuitry enabling a direct time offlight measurement in a CMOS image sensor according to the invention;

FIG. 7 is a chronogram of the control signals of the pixel structure ofFIG. 6, for achieving a sensing phase according to the invention;

FIG. 8 illustrates a variant of the pixel structure of FIG. 6;

FIG. 9 represents another implementation of the amplifier circuit ofFIG. 6;

FIG. 10 highlights the SNR improvement through a high pass or band passfilter to the light pulse width of the light source;

FIG. 11 illustrates a sweeping method to determine an optimum cut-offfrequency of the high pass filter;

FIG. 12 illustrates the efficiency of a pixel operated according to theinvention, when two backscattered light pulses successively reach thepixel closely in time; and

FIG. 13 illustrates the efficiency of a differential digital method forimplementing a high pass filtering in the readout circuitry;

DETAILED DESCRIPTION

An embodiment of a pixel structure in a CMOS imaging sensor for use fordirect time of flight measurement according to the invention isillustrated on FIG. 6, and corresponding signals for controlling thepixel operation are illustrated on FIG. 7.

The pixel structure comprises a photodiode, PHD, as a photodetector. Itis preferably a pinned photodiode. Note that the invention applies aswell to photogates as photodetectors.

It further comprises as in conventional pixel structures of opticsensors, a reset transistor T_(RST) and a transfer transistor T_(TX)connected in series between a voltage supplying node V_(DD)-P, and thephotodetector. The sense node SN is between the reset transistor and thetransfer transistor. It is a capacitive node having a capacitance valueC comprised at least of the equivalent parasitic capacitance at thesense node which is intrinsic to the topology and technology of thepixel structure. In a conventional pixel, the capacitance at the sensenode is mainly determined by the floating diffusion and a high value isdesirable to be able to receive the whole amount of charges having beenintegrated during the integrating period, before the reading phase.

In the pixel structure embodiment of the invention, the capacitance ispreferably reduced at most, which in practice means that it can bereduced to the intrinsic parasitic capacitance of the pixel structure.The capacitance value can then be lower than 5 femtofarads (10⁻¹⁵farads), for example equal to 2 femtofarads. However this is only apreferred condition, which enables to reduce a thermal noise levelbrought into the capacitance by the operation mode of the resettransistor as a nonlinear resistance as has been already explained.Higher capacitance values could be applied.

The sense node provides an output signal of the pixel, which is avoltage signal V_(SN), which is continuously transmitted to a readoutcircuitry 300. The readout circuitry 300 mainly comprises a fast analogto digital convertor ADC and filtering means F to reconstruct in digitalthe waveform of the signal outputted by the sense node with a highsignal to noise ratio at least in a frequency band through the filteringmeans applying at least one of the band pass or high pass filter. Thesignal is then triggered by triggering means TRG to precisely detectpulse locations in the signal. The ADC is a fast ADC, which means thatit operates at a sampling rate SMP that is at least twice 1/τ, where τis the pulse duration of the signal pulses to be detected. For instance,when τ is equal to 5 ns, a minimum SMP rate is 400 Mhz, whichcorresponds to a sampling time of 2.5 ns. However, the lower thesampling time the higher the precision in sensing the pulse location andhence that of the measured distance.

As to the filtering means F, they can be implemented in the digitaldomain, as illustrated in FIG. 6 (after the ADC) which enables toextract further signal information (noise) in a low frequency range. Thefiltering means can either be implemented In the analog domain, and thenthe filtering means F is between a sample and hold circuit S&H and thedigitizing circuit DCV of the ADC. In this case the low frequencyinformation is lost. This will be explained in more details later.

The pixel operation to make a direct time of flight measurementaccording to the invention, is controlled through the control signalsapplied to the reset and transfer transistors of the pixel, and dependson whether the pixel is selected for a DTOF measurement phase, orremains unselected.

When the pixel is unselected, which corresponds to the idle phase ofFIG. 7, the operation corresponds to a conventional reset phase of thepixel:a nominal voltage value (0.2 volts in the example) is applied tothe supply node V_(DD)-P of the pixel through signal V_(DD_RST), andboth the reset transistor and the transfer transistor are switched inthe on state: both RST and TX signals are set to the high (logic)voltage, which enables to reset the sense node SN (through the resettransistor) and the photodiode PHD (through the transfer transistor andthe reset transistor).

When a pixel is selected (SEL signal set to the high logic level—FIG.7):

-   -   the transfer transistor remains in the same ON state, then        operating as a decoupling element between the photodiode node PN        (which has a relatively high capacitance) and the sense node SN,        which minimizes the capacitance at the sense node;    -   the reset transistor is operated in a sub-threshold region        through lowering its gate voltage, through the RST signal, or        increasing its source voltage through the voltage value applied        to the supply node V_(DD)-P of the pixel. In the example the        voltage value applied to the supply node V_(DD)-P goes from a        low value (0.2 volt for instance) up to a high value (3.3 volts,        corresponding to the positive supply voltage of the sensor, for        a given technology), while RST is remained to the ON value (3.3        volts). In another example, the low value (0.2 volts in the        example) can be continuously applied to the supply node V_(DD)-P        of the pixels and the operation as a nonlinear resistance is        then controlled through lowering the gate voltage level through        the RST signal. In practice through one or both of the RST and        V_(DD)-RST signals, it is possible to implement and control the        operation of the reset transistor as a nonlinear resistance in a        very wide range.

As a result of this mode of operation of the selected pixel, a spectraldensity of the signal V_(SN) outputted at the sense node of the pixel issuch that the useful signal information is mainly concentrated in a highfrequency range, and a noise component is mainly concentrated in a lowfrequency range, which is exploited through the post filtering means Fimplemented in the readout circuitry (FIG. 6 or 8) to extract a signalin at least a frequency range concentrating the useful signal with ahigh SNR, before triggering.

In a variant illustrated on FIG. 8, the pixel structure comprises ananti-blooming transistor T_(BG) connected to the photodetector (nodePN), which can be used in the idle state (corresponding to unselectedpixels) to extract the charges from the photodiode, and also from thesense node capacitance through the transfer transistor. Then, forinstance, the reset transistor can be kept off in the idle state andoperated only when the pixel is selected as a nonlinear resistor, in ameasurement phase.

There are other possibilities to implement the nonlinear resistance inthe pixel, other than through a transistor being operated in asub-threshold region, so that the invention applies generally to a pixelstructure comprising a nonlinear resistance in series with the transfertransistor in a selected pixel. However, the use of the reset transistormakes it easy to apply to any known CMOS pixel technology which arebasically comprised of such reset and transfer transistors. That is,only the control circuitry has to be adapted to provide for themeasurement phase as explained.

FIG. 10 illustrates the effect of the post-filtering of the inventionapplied at the side of the readout circuitry applying the fast analog todigital sampling. We assume that in the measurement phase beginning atthe time 0 and lasting one microsecond in the example, a current pulseas the one illustrated on FIG. 3 was generated by the photodectector (asa response to a backscattered light pulse reaching the pixel) in themiddle of the measurement phase, that is at the time 0.5 microsecond inthe figure. The signal V_(SN) is very noisy, which makes the pulsesignal indicated by the arrow indistinguishable from noise. With aproper filtering according to the invention, we obtain the filteredsignal V_(HF): Although the peak of the signal (having the usefulinformation) is reduced, the noise component is greatly attenuated. Thenthe SNR is significantly improved and the location of the pulse can beefficiently detected through triggering means.

That is, although the signal is low pass filtered by the RC circuit ofour CMOS pixel structure, the significant information is not lost andcan be significantly recovered through the post filtering means F.

This remains true even if several pulses appear, corresponding to theecho from several objects in the scene, located at different distances.This is illustrated on FIG. 12. To simplify the explanation, the noiseis omitted. The curve I(t) on top illustrates the photogenerated signalwhen the photodetector receives successively two light pulses P1 and P2.After voltage conversion by the RC circuit at the pixel level in theselected pixel, the corresponding pulses result in two steps ST1 and ST2in the signal V_(SN)(t) that are mixed because of the low pass filterfunction of the RC. However, after our post filtering, both pulses areperfectly recovered in location and amplitude, in the filtered signalV_(HF). In practice we could check that with light pulse duration of 5ns, it was possible to detect close pulses distant from 50 ns only. Thenthe propose invention allows for the detection of the position ofmultiple solid objects appearing as peaks in the signal recovered aftera high pass filtering.

Another aspect of the invention regards the signal transmission pathbetween the pixel and the readout circuit. Indeed, we need to transmitthe signal outputted at the sense node that has high frequencyinformation, in a range of 200 MHZ for example as explained. If theoutput line CL as illustrated on FIG. 6 is used so that several pixelsmay share a same readout circuitry (one selected pixel read at a time),this results in a transmission path having a high resistance andcapacitance between the sense node of the selected pixel and the readoutcircuit. The output line can be modelized by a combination ofcapacitances in parallel and resistances in series with a filteringeffect that may strongly degrade the signal information. According to animproved embodiment of the invention and as illustrated on FIGS. 6 and8: a power amplifier circuit 200 is provided between the sense node SNof the pixel and the output line CL that connects the selected pixel tothe readout circuit 300. The power amplifier has typically a high inputimpedance (so as not to weaken the input signal) and a well controlledoutput impedance to match the impedance of the output line.

Preferably, and as symbolically illustrated in FIG. 6, the poweramplifier 200 combines with an implementation of the output line CL by atransmission line, like a micro strip line MST. Then, the output linehas a relatively well controlled characteristic impedance, whichcorresponds to a distributed R (resistance), L (inductor), C (capacitor)model. This combination of the power amplifier with a transmission lineimproves the fast transmission of the signal to the readout circuit inits whole spectral range (large signal bandwidth transmission).

The power amplifier 200 intrinsically generates noise, in particularshot noise and thermal noise, but it can be kept at a low level at theinput of the amplifier, through setting a driving current in theamplifier higher than the one strictly necessary to have the desiredamplifier function. Higher is the current, lower is the noise, inparticular shot noise and thermal noise. Note that the power amplifierwill also generates flicker noise, which is advantageously removedthrough the band pass or high pass filtering implemented in the readoutcircuit. The power amplifier 200 may have any suitable structure asknown by the man skilled in the art.

In the embodiment of FIG. 6, the amplifier 200 is simply made of asingle stage comprising a pair of transistors T_(FW) and T_(CS) inseries between a nominal supply voltage (Vim) of the sensor and theground, forming a so-called “source follower amplifier”. The transistorT_(cs) is used as a current source, and is biased with an appropriategate voltage V_(bn) to produce the current I_(D) needed to properly fixthe biasing condition of the driving transistor T_(FW) in stronginversion and saturation mode, to operate as a source follower element.As explained supra, to lower the noise at the input of the amplifier thecurrent I_(D) is set higher than the current strictly needed for this.For instance, while a current I_(D) around 22·10⁻⁶ amperes is enough tohave the source follower operation, it will be set around 90·10⁻⁶amperes to lower the induced noise at the sense node to an acceptablelevel.

An advantage of such an embodiment of the amplifier is that it is easilyimplemented inside the pixel structure, and requires very low area inthe pixel (only two transistors). Then the fill factor quantumefficiency (FFQE) of the pixel is not affected. However the impedanceand power gain characteristics of such a power amplifier are adequateonly when the intrinsic resistance and capacitance of the output lineare low, which means in practice that only one or a few pixels (arrangedin a same column) are connected to a same readout circuitry 300 througha same output line CL. In case of a high capacitive output line, it isdesirable in practice to design or choose a power amplifier 200 having amore complex structure, with several successive amplifier stages, forimproving the signal characteristics further. Then further noisereduction and/or power gain are obtained. Also, it relaxes theresolution constraint of the analog to digital convertor throughenhancing the SNR.

FIG. 9 illustrates an embodiment of such an improved amplifier for thesake of example only. In this example, the power amplifier 200 includesa first amplifier stage based on transistors of both N and P type (CMOS)with a mirror current arrangement, followed by a source follower stage(T_(FW), T_(CS)). The invention is not limited to a particularembodiment of the power amplifier 200 and the man skilled in the artwill take into account the different application constraints (surface,impedances involved, . . . ) to design or choose the amplifier structurefitted for a DTOF CMOS sensor according to the invention.

It should be noted that in some cases the power amplifier 200 is notneeded. In particular, if all pixels in a sensor must continuously besensed, which means that they are all selected simultaneously to applythe measurement phase illustrated on FIG. 7, then each pixel must haveits own readout circuit 300 comprising at least the fast analog todigital convertor, and possibly the post-filtering means whenimplemented in analogic at least. In this case the output sense node candirectly be connected to its respective readout circuitry which can bedone inside the pixel, or implemented on a separate substrate which isstacked with that of the sensor (3D stacking method).

Regarding the filtering means F, as indicated above, they can operateeither in the analog (FIG. 8) or the digital domain (FIG. 6). In bothcases, the filtering means are designed in account of the high frequencyrange band which according to the invention contains most of the signalcomponent, while most of the noise component has been filtered out bythe RC circuit operation at the level of the pixel.

In an embodiment, the post-filtering means implements a high pass filtercomplementary to the low pass filter (RC circuit within the pixelstructure). This can be achieved very simply through a differentiatingtechnic making the difference between two successive samples of signal,namely through subtracting from the current sample, the previous sample.This is illustrated on FIG. 13: the first curve on top illustrates thecontinuous voltage signal V_(SN)(t) outputted at the sense node. Thearrow indicates a location in time of a pulse of signal to be detectedwhich is undistinguishable from noise. The second curve in the middle isthe reconstructed digitized waveform signal V_(SMP)(t) outputted by thefast analog to digital convertor 300, before filtering. Then the thirdcurve is the differentiated sampled signal obtained through thedifferentiating method where the filtered sample value at the nthsampling time is equal to the sample value at the nth sampling time lessthe sample value at the n−1th sampling time. And this result in a signalpeak that can be detected from the noise. This subtraction is very easyto implement in digital but in analogic as well.

More complex filters can be implemented to improve the SNR. Inparticular a band pass filter may be specially fitted to optimize thesignal to noise ratio, based on the spectral characteristics of thesignal obtained at the sense node. Typically, for light pulses of σ=5nanoseconds, we have explained that the useful signal at the sense nodeis mainly concentrated in a frequency range around 200 Mhz (=1/σ). It ispossible in a given topology to estimate this frequency range in whichthe signal is mainly concentrated, for instance 200 Mhz±10 Mhz, and toimplement a corresponding band-pass filter.

Also a cutoff frequency of the high pass filter to implement can beevaluated for a given sensor and a given application through sweepingthe frequency to find the one that optimizes the SNR. This isillustrated by FIG. 11.

We may then prefer implementing the post-processing filter in thedigital domain, that is after the ADC (FIG. 6), which generally means ina processing device, outside the sensor. Then we can implement afiltering process (computation) that combines a high filtering functionto recover a pulse position information, and a low filtering function toextract additional signal information including background light signalinformation. Such background information enables to improve thereliability of the measure.

A practical implementation of the post-processing will depend on thecontext/application. In particular, if the application if perfectlyknown, then the post-processing filter can be fixed in analog into thesensor (FIG. 8) and the overall system will be cheaper, with lessexternal components. This is possible because the fundamental filteringoperation to apply according to the invention is very simple and easy toimplement in hardware as has been explained with reference to FIG. 13(subtraction between two successive samples). Now if the filter is fixedinto the sensor, flexibility of the product is lost. Where improved SNRis desirable, a digital implementation is preferable.

1. A CMOS imaging sensor for detecting time occurrence of light pulseshaving a given pulse duration_(τ), comprising: pixels, each pixel (P)having a pixel structure comprising at least: a photodetector (PhD)operating as a current source, a transfer transistor (T_(TX)) connectedin series between a sense node (SN) and the photodetector, a nonlinearresistor (R), not included in the transfer transistor (T_(TX)),connected and operated continuously between a voltage supplying node(V_(DD-P)) of the pixel and the sense node (SN), the sense node (SN)being a capacitive sense node, a control circuit (100) for controlling ameasurement phase in at least a selected pixel (P) for measuring a timeoccurrence of a light pulse reaching the photodetector in the pixel,wherein in the selected pixel a voltage reference is applied to thevoltage supplying node, and the transfer transistor is the ON state allalong the measurement phase, the transfer transistor then operating as adecoupling element between a photodetector node (PN) and the sense node(SN), and the nonlinear resistor combining with a capacitance (C) at thesense node form a RC circuit that has a low pass filtering function andhigh frequency integration function that produces a voltage signal(V_(SN)) at the sense node that has a main signal component comprisingat least a pulse location information in a high frequency range, and anoise component mainly concentrated in a low frequency range spacedapart from said high frequency range, and a readout circuitry of avoltage signal from the sense node of a selected pixel, which comprisesat least: an analog to digital converter (300) applying a high samplingtime in respect of the pulse width duration, and a filter (F) configuredto apply continuously at least one of a band pass or high pass filter,before or after analog to digital conversion, having the effect ofincreasing the signal to noise ratio in at least a frequency band aroundthe main signal component.
 2. The CMOS imaging sensor of claim 1,wherein the nonlinear resistor is a reset transistor (T_(RS)), differentfrom the transfer transistor (T_(TX)), operated in a sub-thresholdregion in a selected pixel, which transistor is further used as a switchand turned on to maintain the sense node (SN) at a voltage reference innon-selected pixels.
 3. The CMOS imaging sensor of claim 2, wherein thecontrol circuit is configured to operate said transistor as a nonlinearresistance in a selected pixel through controlling at least one of agate signal (RST) or a voltage reference (V_(DD_RST)) applied to thevoltage supplying node of the pixel.
 4. The CMOS imaging sensor of claim2, wherein the transfer transistor is also set in the ON mode in theunselected pixels.
 5. The CMOS imaging sensor of claim 1, in which thecapacitance at the sense node of a pixel is no more than 5 femtofarads.6. The CMOS imaging sensor of claim 1, in which the filter (F) isimplemented in the analog to digital convertor, after a sample and holdcircuit (301) and before a digitizing circuit (302).
 7. The CMOS imagingsensor of claim 1, in which the filter (F) is implemented in digital andoperates on the digitized samples provided by the analog to digitalconvertor (300).
 8. The CMOS imaging sensor of claim 7, wherein digitalimplementation achieves a high pass filter through computing differencesbetween two successive sampled signals.
 9. The CMOS imaging sensor ofclaim 7, in which the filter is a digital filter which is configured toextract pulse location and amplitude of the signal information in a highfrequency range and at least background light noise in a low frequencyrange.
 10. The CMOS imaging sensor of claim 1, further comprising apower amplifier (200) between a sense node (SN) of a selected pixel andthe readout circuit (300), the power amplifier having a high impedanceinput and an impedance output matching an impedance of an output line(CL) to the readout circuit.
 11. The CMOS imaging sensor of claim 10, inwhich the power amplifier (200) comprises at least a source followeroutput stage (T_(FW), T_(LD)).
 12. The CMOS imaging sensor of claim 10,in which the output line is a transmission line (MST).
 13. A direct timeof flight system comprising a CMOS imaging sensor as claimed in claim 1,as a pulse light receiver, in which a light pulse source is operated toemit only one or a reduced number of light pulses, in a number lowerthan or equal to 5, per measurement phase.
 14. A method to operate apixel in a CMOS imaging sensor for a measure of time through triggeringa pulse location in a voltage signal waveform corresponding to a lightpulse having reached the pixel, the pixel having a structure comprisinga photodetector (PhD) exposed to light and operating as a currentsource, a first reset transistor (T_(RST)) connected between a supplyvoltage reference source and a capacitive sense node (SN), wherein thesense node outputs a voltage signal, a second transfer transistor(T_(TX)) connected in series between the sense node (SN) and thephotodetector, wherein the method applies a measurement phase comprisingcontinuously operating the first reset transistor as a nonlinearresistance, the second transfer transistor being turned on, continuouslyfiltering out the voltage signal through at least one of a band pass ora high pass filtering, resulting in increasing the signal to noise ratioin the frequency range of the filtering, analog to digital conversionprior or after said continuous filtering, at a sampling frequency (SMP)higher than a frequency value corresponding to the pulse light duration(τ), determining pulse location(s) in the filtered digital signalthrough level triggering.
 15. The CMOS imaging sensor of claim 3,wherein the transfer transistor is also set in the ON mode in theunselected pixels.
 16. The CMOS imaging sensor of claim 4, in which thecapacitance at the sense node of a pixel is no more than 5 femtofarads.17. The CMOS imaging sensor of claim 5, in which the filter (F) isimplemented in the analog to digital convertor, after a sample and holdcircuit (301) and before a digitizing circuit (302).
 18. The CMOSimaging sensor of claim 5, in which the filter (F) is implemented indigital and operates on the digitized samples provided by the analog todigital convertor (300).
 19. The CMOS imaging sensor of claim 9, furthercomprising a power amplifier (200) between a sense node (SN) of aselected pixel and the readout circuit (300), the power amplifier havinga high impedance input and an impedance output matching an impedance ofan output line (CL) to the readout circuit.